Abstract
This paper discusses a framework for algorithm-architecture synergy for (1) performance evaluation and (2) FPGA implementation complexity analysis of linear massive MIMO detection techniques. Three low complexity implementation techniques of the zero-forcing (ZF) based linear detection are evaluated, namely, Neumann series expansion (NSE), Gauss–Seidel (GS) and a proposed recursive Gram matrix inversion update (RGMIU) techniques. The performance analysis framework is based on software-defined radio platform. By extrapolating the real data measured average error vector magnitude versus a number of served single-antenna user terminals, GS and RGMIU are showing no performance degradation with respect to ZF with direct matrix inversion. It is shown that under high load regime NSE and GS require more processing iterations at the expense of increased processing latency. We, therefore, consider a unified approach for field-programmable gate array based implementation complexity analysis and discuss the required baseband processing resources for real-time transmission. Due to the wide differences of NSE, GS and RGMIU in terms of performance, processing complexity and latency, practical deployment and real-time implementation insights are derived.
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Notes
Note that these limits are also establish by 3GPP for LTE transmitter’s modulation accuracy performance.
We also expect early deployments will be limited to less than 64 antenna elements for the sake of size and energy efficiency..
In fact, the LTE uplink uses SC-FDMA whereas the downlink adopts FDMA. This is mainly due to the fact that handsets can not afford to support a high peak to average power ratio (PAPR) inherent in FDMA. Our reference SDR design uses FDMA on both links to keep it symmetrical and ease waveform development especially in assigning uplink pilots to the users. Since the processing is done after IFFT, we do not expect noticeable impact unless the radio at the UE side is running at high output power which would make PAPR affect the uplink performance. Particularly, the performance would be decreased in the UL for UEs operating at high power, e.g. signaling at the cell margin. We (coarse) tuned the output power so that the UEs operate in a safe non-saturated region especially that the set-up is constrained to line-of-sight.
For instance, one can alter the transceiver’s local oscillator phase noise by tuning the charge-bump current.
Over the air synchronization and coarse frequency and sampling rate offset compensation is consider for future work.
(24 bits/subcarrier × 1200 subcarrier/symbol × 5 symbols/sub-frame)/(1 ms/sub-frame).
(6 bits/subcarrier × 1200 subcarriers/symbol × 5 symbols)/1 ms.
This expression is accurate if the number of UTs \( K \cong 2^{N} \) where \( N \) is a positive integer.
The BRAM resources are not shown here due to their low use ratio.
Out of the scope of this work, it is interesting to investigate how one can leverage on weights interpolation instead of computing them on every subcarrier.
576 DSP48s are required for 12 × 12 DWMC core.
The non-pipelined DSP48s maximum frequency is 260 MHz and 335 MHz for low and high speed grade Xilinx’s Kintex UltraScale XCKU115 FPGA.
For the sake of clarification; based on the DWMC core, data and processing flow dependencies determine the overall latency which explains why NSE (order 4) has lower latency compared to ZF (based on Cholesky decomposition).
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This work was supported in part by NUTAQ innovation and the Natural Sciences and Engineering Research Council of Canada.
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Appendix
Appendix
To make the paper self-contained this appendix derives the matrix inversion of a large \( K \times K \) matrix of the form \( {\mathbf{H}}^{H} {\mathbf{H}} \). We propose to apply the matrix inversion update lemma when a new column is added [16]. Assume we have the inverse of a \( \left( {K - 1} \right) \times \left( {K - 1} \right) \) matrix \( {\varvec{\Delta}}_{K - 1}^{ - 1} = {\mathbf{H}}_{1:K - 1}^{H} {\mathbf{H}}_{1:K - 1}^{{}} \) (note that we have adopted the Matlab index notation \( 1:K - 1 \) in \( {\mathbf{H}}_{1:K - 1}^{{}} \) to designate columns 1 to \( K - 1 \) of \( {\mathbf{H}} \)). Therefore, the inverse of a \( K \times K \) matrix \( {\varvec{\Delta}}_{K}^{ - 1} = {\mathbf{H}}_{1:K}^{H} {\mathbf{H}}_{1:K}^{{}} \) can be computed as follow
where \( c = \frac{1}{{\left( {{\mathbf{H}}_{K:K}^{H} {\mathbf{H}}_{K:K}^{{}} } \right) - \left( {{\mathbf{H}}_{1:K - 1}^{H} {\mathbf{H}}_{K:K}^{{}} } \right)^{H} {\varvec{\Delta}}_{K - 1}^{{}} \left( {{\mathbf{H}}_{1:K - 1}^{H} {\mathbf{H}}_{K:K}^{{}} } \right)}} \) and \( {\varvec{\Gamma}} = {\varvec{\Delta}}_{K - 1}^{{}} + c{\kern 1pt} {\varvec{\Delta}}_{K - 1}^{{}} \left( {{\mathbf{H}}_{K:K}^{H} {\mathbf{H}}_{1:K - 1}^{{}} } \right)^{H} \left( {{\mathbf{H}}_{K:K}^{H} {\mathbf{H}}_{1:K - 1}^{{}} } \right){\varvec{\Delta}}_{K - 1}^{H} \).
If we set \( {\mathbf{z}} \triangleq {\mathbf{H}}_{K:K}^{{}} \), \( {\mathbf{y}}_{1} \triangleq {\mathbf{H}}_{1:K - 1}^{H} {\mathbf{z}} \),\( {\mathbf{y}}_{2} \triangleq {\varvec{\Delta}}_{K - 1}^{{}} {\mathbf{y}}_{1} \), and \( {\mathbf{y}}_{3} \triangleq c{\kern 1pt} {\kern 1pt} {\mathbf{y}}_{2} \) then \( c = {1 \mathord{\left/ {\vphantom {1 {\left( {{\mathbf{z}}^{H} {\mathbf{z}} - {\mathbf{y}}_{1}^{H} {\mathbf{y}}_{2}^{{}} } \right)}}} \right. \kern-0pt} {\left( {{\mathbf{z}}^{H} {\mathbf{z}} - {\mathbf{y}}_{1}^{H} {\mathbf{y}}_{2}^{{}} } \right)}} \) and \( {\varvec{\Gamma}} = {\varvec{\Delta}}_{K - 1}^{{}} + c{\kern 1pt} {\mathbf{y}}_{2}^{{}} {\mathbf{y}}_{2}^{H} \).
Applying Eq. (6) successively from the second column all the way to the last column \( K \), the algorithm, dubbed recursive gram matrix inversion update (RGMIU), is outlined in Table 1.
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Ahmed Ouameur, M., Massicotte, D., Akhtar, A.M. et al. Performance evaluation and implementation complexity analysis framework for ZF based linear massive MIMO detection. Wireless Netw 26, 4079–4093 (2020). https://doi.org/10.1007/s11276-020-02318-y
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DOI: https://doi.org/10.1007/s11276-020-02318-y