Abstract
This paper studies differential space-time modulation using diversity-encoded differential amplitude and phase shift keying (DAPSK) for the multiple-input multiple-output (MIMO) system over independent but not identically distributed (inid) time-correlated Rician fading channels. An asymptotic maximum likelihood (AML) receiver is developed for differentially detecting diversity-encoded DAPSK symbol signals by operating on two consecutive received symbol blocks sequentially. Based on Beaulieu’s convergent series, the bit error probability (BEP) upper bound is analyzed for the AML receiver over inid time-correlated Rician fading channels. Particularly, an approximate BEP upper bound of the AML receiver is also derived for inid time-invariant Rayleigh fading channels with large received signal-to-noise power ratios. By virtue of this approximate bound, a design criterion is developed to determine the appropriate diversity encoding coefficients for the proposed DAPSK MIMO system. Numerical and simulation results show that the AML receiver for diversity-encoded DAPSK is nearly optimum when the average received signal-to-noise power ratios are high and the channel is heavily correlated fading and can provide better error performance than conventional noncoherent MIMO systems when the effect of non-ideal transmit power amplification is taken into account.
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Notes
Notably, all entries of the transmitted symbol matrix for DUSTM using APSK-based OSTCC take value in irregular symbol constellations. However, its PAPR value is fixed by 3 dB [16].
As indicated in [23], the error performance of DAPSK is highly sensitive to the amplitude ring ratio \(\mu \) in single-input multiple-output antenna systems, especially when the number of amplitude rings is large. This amplitude ring ratio parameter will be optimized in Sect. 4 through minimizing the BEP upper bound developed therein.
The rows of Walsh Hadamard and discrete Fourier transform matrices can be used as the dispersion vectors in the paper.
When \(\mathcal {S}\) is given, \(|x_{k}^{(i-1)}|\) and \( d_{k}=x_{k}^{(i)}/x_{k}^{(i-1)}\) that are required in the following analysis are, respectively given by \(|x_{k}^{(i-1)}|\, =\lambda \mu ^{(\beta _{k}^{(a)}a^{(i-1)}+a_{k}^{(0)})_{N}}\) and \(d_{k}=\mu ^{(\beta _{k}^{(a)}(a^{(i-1)}+\varDelta a)_{N}+a_{k}^{(0)})_{N}-(\beta _{k}^{(a)}a^{(i-1)}+a_{k}^{(0)})_{N}}\exp \{\frac{j2\pi \beta _{k}^{(p)}\varDelta b}{M}\}\) provided with predetermined \(\beta _{k}^{(a)},\, \beta _{k}^{(p)}\), and \(a_{k}^{(0)}\) for \(k\in \mathcal {Z}_{L_{t}}^{+}\).
As shown in (11), \(|d_{k}|\, =\mu ^{(\beta _{k}^{(a)}(a^{(i-1)}+\varDelta a)_{N}+a_{k}^{(0)})_{N}-(\beta _{k}^{(a)}a^{(i-1)}+a_{k}^{(0)})_{N}}\) takes the form \(\mu ^{0}\) when \( \varDelta a=0\). When \(\varDelta a\ne 0\), it takes the form \(\mu ^{(\beta _{k}^{(a)}\varDelta a)_{N}}\) if \(|d_{k}|\, >1\) and \(\mu ^{(\beta _{k}^{(a)}\varDelta a)_{N}-N}\) if \(|d_{k}|\, <1\).
For conventional DNUSTM and DUSTM techniques, each data symbol is differentially encoded into two adjacently transmitted \(L_{t}\times L_{t}\) matrices and each transmitted matrix is emitted every \(L_{t}T_{s}\) seconds. Because \(L_{t}\) encoded symbols are transmitted at each antenna in each \( L_{t}T_{s}\) seconds, all the existing techniques operating at a rate of \(R\) bits per transmit antenna per \(L_{t}T_{s}\) provide a bandwidth efficiency of \(R\) bits\(/\)sec\(/\)channel use.
In the following, the initial transmitted symbol matrices used for all compared DNUSTM and DUSTM techniques are set to the identity matrix.
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Appendix
Appendix
1.1 Derivation of (18)
Given \(\mathcal {S}\) and \(\mathcal {S}_{d},\, Q_{kl}\) in (17) is a conditionally GQS of \(U_{kl}\) and \(V_{kl}\) which have means \(\overline{U} _{kl}\triangleq \mathbb {E}\{U_{kl}|\mathcal {S},\mathcal {S}_{d}\}=\sqrt{ \gamma _{kl}/\varGamma _{kl}}\tau _{kl}(d_{k}-\widehat{d}_{k})x_{k}^{(i-1)}\) and \(\overline{V}_{kl}\triangleq \mathbb {E}\left\{ V_{kl}|\mathcal {S}, \mathcal {S}_{d}\right\} =0\) and second central moments
With reference to [27, eq. (B-5)], the conditional MGF \(\varPhi _{Q_{kl}}(s|\mathcal {S},\mathcal {S}_{d})\) is given by (18) with
Note that \(v_{kl}^{(1)}v_{kl}^{(2)}=-u_{kl}\). Using (25)–(27 ), \(\phi _{kl}^{(1)}\phi _{kl}^{(2)}-|\phi _{kl}^{(3)}|^{2}\) can be further expressed as
For \(d_{k}\ne \widehat{d}_{k},\, \phi _{kl}^{(1)}\phi _{kl}^{(2)}-|\phi _{kl}^{(3)}|^{2}\) is always positive, and then \(v_{kl}^{(1)}>0\) and \( v_{kl}^{(2)}<0\) from (28).
1.2 Derivations of (23) and (24)
When \(\varOmega _{kl}=0\) and \(\rho _{kl}=1\) for all \(k\in \mathcal {Z} _{L_{t}}^{+}\) and \(l\in \mathcal {Z}_{L_{r}}^{+}\), the conditional MGF \(\varPhi _{Q}(s|\mathcal {S},\mathcal {S}_{d})\) in (18) specializes to
where \(v_{kl}\) and \(u_{kl}\) in (29) simplify, respectively to
When \(\gamma _{t}\gg 1,\, v_{kl}\) and \(u_{kl}\) can be further approximated as \(v_{kl}\approx 1/2\) and \(u_{kl}\approx (1+|\widehat{d}_{k}|^{2})/(|d_{k}- \widehat{d}_{k}|^{2}|x_{k}^{(i-1)}|^{2}g_{k}\varGamma _{kl}\gamma _{t})\), and thus \(v_{kl}^{(1)}\) and \(v_{kl}^{(2)}\) in (28) are approximated as \( v_{kl}^{(1)}\approx 0\) and \(v_{kl}^{(2)}\approx -1\) for \(k\in \mathcal {Z} _{L_{t}}^{+}\) and \(l\in \mathcal {Z}_{L_{r}}^{+}\) because \(u_{kl}\) is much smaller than \(v_{kl}\) for large \(\gamma _{t}\) values. Using this approximation, \(\varPhi _{Q}\left( s|\mathcal {S},\mathcal {S}_{d}\right) \) in ( 31) can be approximated for \(\gamma _{t}\gg 1\) as
Putting the high SNR approximation of \(\Pr \{Q<\vartheta |\mathcal {S}, \mathcal {S}_{d}\}\) in (32) into (21) yields
Using Leibniz’s rule of differentiation in [28, eq. (0.42)], we have
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Huang, CH., Chung, CD. Differential Space-Time Modulation Using DAPSK Over Rician Fading Channels. Wireless Pers Commun 78, 1021–1046 (2014). https://doi.org/10.1007/s11277-014-1799-7
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DOI: https://doi.org/10.1007/s11277-014-1799-7